Systems and methods for calibrating power measurements in an electrosurgical generator

ABSTRACT

The disclosed electrosurgical systems and methods accurately determine the power actually applied to a load by using equalizers to calibrate the power measurements. The electrosurgical systems include an electrosurgical generator and an electrosurgical instrument coupled to the electrosurgical generator through an electrosurgical cable. The electrosurgical generator includes an electrical energy source, voltage and current detectors, equalizers that estimate the voltage and current applied to a load, and a power calculation unit that calculates estimated power based upon the estimated voltage and current. The methods of calibrating an electrosurgical generator involve applying a resistive element across output terminals of the electrosurgical generator, applying a test signal to the resistive element, measuring the magnitude and phase angle of voltage and current components of the test signal within the electrosurgical generator, estimating the magnitude and phase angle of the voltage and current at the resistive element using equalizers, and determining gain correction factors and minimum phase angles for the equalizers.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a divisional of U.S. application Ser. No. 13/085,278filed on Apr. 12, 2011, now U.S. Pat. No. 8,968,293, the entire contentof which is incorporated by reference in this application.

BACKGROUND

1. Technical Field

The present disclosure generally relates to electrosurgery. Moreparticularly, the present disclosure relates to systems and methods forcalibrating power measurements within an electrosurgical generator.

2. Background of Related Art

Electrosurgery involves the application of high-frequency electriccurrent to cut or modify biological tissue during a surgical procedure.Electrosurgery is performed using an electrosurgical generator, anactive electrode, and a return electrode. The electrosurgical generator(also referred to as a power supply or waveform generator) generates analternating current (AC), which is applied to a patient's tissue throughthe active electrode and is returned to the electrosurgical generatorthrough the return electrode. The alternating current typically has afrequency above 100 kilohertz to avoid muscle and/or nerve stimulation.

During electrosurgery, the alternating current generated by theelectrosurgical generator is conducted through tissue disposed betweenthe active and return electrodes. The tissue's impedance converts theelectrical energy (also referred to as electrosurgical energy)associated with the alternating current into heat, which causes thetissue temperature to rise. The electrosurgical generator controls theheating of the tissue, by controlling the electric power (i.e.,electrical energy per time) provided to the tissue. Although many othervariables affect the total heating of the tissue, increased currentdensity usually leads to increased heating. The electrosurgical energyis typically used for cutting, dissecting, ablating, coagulating, and/orsealing tissue.

The two basic types of electrosurgery employed are monopolar and bipolarelectrosurgery. Both of these types of electrosurgery use an activeelectrode and a return electrode. In bipolar electrosurgery, thesurgical instrument includes an active electrode and a return electrodeon the same instrument or in very close proximity to one another,usually causing current to flow through a small amount of tissue. Inmonopolar electrosurgery, the return electrode is located elsewhere onthe patient's body and is typically not a part of the electrosurgicalinstrument itself. In monopolar electrosurgery, the return electrode ispart of a device usually referred to as a return pad.

An electrosurgical generator includes a controller that controls thepower applied to a load, i.e., the tissue, over some period of time. Thepower applied to the load is controlled based upon the power determinedat the output of the electrosurgical generator and a power level set bythe user or a power level needed to achieve a desired tissue effect. Thepower at the output of the electrosurgical generator is determined bymeasuring the voltage and current at the output of the electrosurgicalgenerator and calculating the average power based upon the measuredvoltage and current.

The voltage and current measured by the sensors at the output of theelectrosurgical generator, however, may not equal the actual voltage andcurrent applied to the load, i.e., the tissue, because of errors in thevoltage and current measurements. These measurement errors may be causedby parasitics in the cable connecting the electrosurgical generator tothe electrosurgical instrument, parasitics in the analog processingcircuitry, and/or delays of the analog to digital conversion process. Asa result, the power calculations may be inaccurate and may lead toimproper control of the electrosurgical energy applied to the tissue.

SUMMARY

The system and method of the present disclosure accurately determinesthe power actually applied to tissue by calibrating the powermeasurements within an electrosurgical generator using equalizers at adesired frequency or over a narrow bandwidth of frequencies. Theequalizers have low computational complexity and may be implementedusing commonly available microprocessors, field programmable gate arrays(FPGAs), or digital signal processors (DSPs).

In one aspect, the present disclosure features an electrosurgicalgenerator system. This system includes an electrosurgical generator andan electrosurgical instrument coupled to the electrosurgical generatorthrough an electrosurgical cable. The electrosurgical instrument isconfigured to apply electrosurgical energy to body tissue. Theelectrosurgical generator includes an radio frequency (RF) electricalenergy source, a voltage detector coupled to the RF electrical energysource, a current detector coupled to the RF electrical energy source,an equalizer unit configured to equalize the voltage detected by thevoltage detector and the current detected by the current detector, and apower calculation unit that calculates power based upon the equalizedvoltage and current.

In some embodiments, the electrosurgical generator includes a digitalsignal processor (DSP), which includes the equalizer unit and the powercalculation unit. The equalizer unit may include a Least Mean Squares(LMS) adaptive filter, a gain and a fractional delay line, at least onegain and an all-pass delay filter, or a bandpass parametric equalizer.The bandpass parametric equalizer may be a shelving boost filter, ashelving cut filter, or a peak filter. The equalizer unit may alsoinclude a polyphase filter and decimator configured to performequalization, filtering, and decimation as a combined function.

In some embodiments, the electrosurgical generator system furtherincludes analog-to-digital converters electrically coupled to thecurrent and voltage detectors. The power calculation unit calculatesactual power applied to the electrosurgical instrument.

The present disclosure, in another aspect, features a method ofcontrolling an electrosurgical generator system. The method includesgenerating RF electrical energy, sensing the voltage and current of theRF electrical energy, equalizing the voltage and current of the RFelectrical energy, calculating power based upon the equalized voltageand current, and modifying the power of the RF electrical energy basedupon the calculated power to achieve desired tissue effects.

In some embodiments, equalizing the voltage and current of the RFelectrical energy includes filtering the sensed voltage and the sensedcurrent with an LMS adaptive filter. In other embodiments, equalizingthe sensed voltage and the sensed current of the RF electrical energyincludes applying a gain to the sensed voltage and the sensed currentand delaying the result of applying a gain to the sensed voltage and thesensed current to correct unequal group delay. The delay may be afractional delay line filter. In yet other embodiments, equalizing thesensed voltage and the sensed current of the RF electrical energyincludes applying a gain to the sensed voltage and the sensed currentand filtering the result with an all-pass delay filter. In yet otherembodiments, equalizing the sensed voltage and the sensed current of theRF electrical energy includes equalizing the sensed voltage and thesensed current of the RF electrical energy using a bandpass parametricequalizer, such as a shelving boost filter, a shelving cut filter, or apeak filter.

In some embodiments, the method includes converting the sensed voltageand current to digital form. Also, calculating power based upon theequalized voltage and current includes calculating the actual averagepower applied to a load. In addition, modifying the power of the RFelectrical energy includes comparing the calculated power to a presetpower value or desired power value based on the calculated tissueimpedance, and modifying the power of the electrosurgical energy basedupon the result of comparing the calculated power to the preset powervalue or desired power value.

The present disclosure, in yet another aspect, features a method ofcalibrating power measurements in an electrosurgical generator. Themethod includes selecting a resistive element; applying the resistiveelement across the output terminals of the electrosurgical generator;generating a test signal at a desired frequency; applying the testsignal to the resistive element; measuring first magnitude values andfirst phase angle values of voltage and current components of the testsignal at the output terminals; estimating second magnitude values andsecond phase angle values for the voltage and current components of thetest signal using a first equalizer for the voltage component and asecond equalizer for the current component; determining gain correctionfactors for the first and second equalizers based on the measured andestimated magnitudes of the voltage and current components of the testsignal; and determining the minimum phase angle of the first and secondequalizers based on the measured and estimated phase angles of thevoltage and current components of the test signal.

BRIEF DESCRIPTION OF THE DRAWINGS

Various embodiments of the present disclosure are described withreference to the accompanying drawings wherein:

FIG. 1 is an illustration of an electrosurgical system in accordancewith embodiments of the present disclosure;

FIG. 2 is a block diagram of the electrosurgical generator of FIG. 1coupled to a medical instrument in accordance with embodiments of thepresent disclosure;

FIG. 3 is a block diagram of the equalizer of FIG. 2 in accordance withan embodiment of the present disclosure;

FIG. 4 is a block diagram of the equalizer of FIG. 2 in accordance withanother embodiment of the present disclosure;

FIG. 5 is a block diagram of the equalizer of FIG. 2 that includes afractional delay line filter block in accordance with another embodimentof the present disclosure;

FIGS. 6A and 6B are block diagrams of the fractional delay line block ofFIG. 5 in accordance with embodiments of the present disclosure;

FIGS. 7A-7C are block diagrams of equalizers in accordance with otherembodiments of the present disclosure;

FIG. 8 is a block diagram of an equalizer in the form of a digital combfilter in accordance with other embodiments of the present disclosure;

FIG. 9 is a block diagram of the electrosurgical generator of FIG. 1coupled to a test accessory in accordance with embodiments of thepresent disclosure; and

FIG. 10 is a flow diagram of a method of calibrating power measurementsin an electrosurgical generator in accordance with other embodiments ofthe present disclosure.

DETAILED DESCRIPTION

FIG. 1 illustrates a bipolar and monopolar electrosurgical system 100 inaccordance with embodiments of the present disclosure. Theelectrosurgical system 100 includes an electrosurgical generator 102that measures and calculates the power delivered to a load through anelectrosurgical instrument. The electrosurgical generator 102 performsmonopolar and bipolar electrosurgical procedures, including vesselsealing procedures. The electrosurgical generator 102 may include aplurality of outputs (e.g., terminals 104 and 106) for interfacing withvarious electrosurgical instruments (e.g., a monopolar active electrode108, a return pad 110, bipolar electrosurgical forceps 112, and afootswitch (not shown)). The electrosurgical generator 102 also includeselectronic circuitry that generates radio frequency power for variouselectrosurgical modes (e.g., cutting, coagulating, or ablating) andprocedures (e.g., monopolar, bipolar, or vessel sealing).

The electrosurgical system 100 includes a monopolar electrosurgicalinstrument 114 having one or more electrodes 108 for treating tissue ofa patient (e.g., an electrosurgical cutting probe or ablationelectrodes). Electrosurgical energy, e.g., radio frequency (RF) current,is supplied to the instrument 114 by the electrosurgical generator 102via a supply line 116, which is connected to an active terminal 104 ofthe electrosurgical generator 102, allowing the instrument 114 tocoagulate, seal, ablate and/or otherwise treat tissue. Theelectrosurgical current returns from the tissue via a return line 118 ofthe return pad 110 to a return terminal 106 of the electrosurgicalgenerator 102. The active terminal 104 and the return terminal 106 mayinclude connectors (not explicitly shown) configured to interface withplugs (also not explicitly shown) disposed at the end of the supply line116 of the instrument 114 and at the end of the return line 118 of thereturn pad 110.

The electrosurgical system 100 includes return electrodes 120 and 122within return pad 110 that are arranged to minimize the risk of tissuedamage by maximizing the overall contact area with the patient's tissue.In addition, the electrosurgical generator 102 and the return pad 110may be configured to monitor tissue-to-patient contact to insure thatsufficient contact exists between the return pad 110 and the patient tominimize the risk of tissue damage.

The electrosurgical system 100 also includes a bipolar electrosurgicalforceps instrument 112 having two or more electrodes (e.g., electrodes124, 126) for treating tissue of a patient. The instrument 112 includesopposing jaw members 134, 136. The first jaw member 134 includes anactive electrode 124 and the second jaw member 136 includes a returnelectrode 126. The active electrode 124 and the return electrode 126 areconnectable to the electrosurgical generator 102 through cable 128,which includes a supply line 130 and a return line 132. The supply line130 is connectable to the active terminal 104 and the return line 132 isconnectable to the return terminal 106. The instrument 112 connects tothe active terminal 104 and the return terminal 106 of theelectrosurgical generator 102 through a plug (not explicitly shown)disposed at the end of the cable 128.

The electrosurgical generator 102 may be any suitable type of generator(e.g., electrosurgical or microwave) and may include a plurality ofconnectors to accommodate various types of electrosurgical instruments(e.g., instrument 114 and electrosurgical forceps 112). Theelectrosurgical generator 102 may also be configured to operate in avariety of modes, such as ablation, monopolar cutting, bipolarcoagulation, and other modes. The electrosurgical generator 102 mayinclude a switching mechanism (e.g., relays) to switch the supply of RFenergy between the connectors. For example, when the instrument 114 isconnected to the electrosurgical generator 102, the switching mechanismswitches the supply of RF energy to only the monopolar plug. The activeterminal 104 and the return terminal 106 may be coupled to a pluralityof connectors (e.g., inputs and outputs) of the electrosurgicalgenerator 102 to power a variety of instruments.

The electrosurgical generator 102 includes suitable input controls(e.g., buttons, activators, switches, or touch screens) for controllingthe electrosurgical generator 102. In addition, the electrosurgicalgenerator 102 may include one or more display screens for providing theuser with a variety of output information (e.g., intensity settings andtreatment complete indicators). The controls allow the user to adjustparameters of the RF electrical energy (e.g., the power or the waveform)so that they are suitable for a particular task (e.g., coagulating,tissue sealing, or cutting). The instruments 112 and 114 may alsoinclude a plurality of input controls that may be redundant with certaininput controls of the electrosurgical generator 102. Placing the inputcontrols at the instruments 112 and 114 allow for easier and fastermodification of RF energy parameters during the surgical procedurewithout requiring interaction with the electrosurgical generator 102.

FIG. 2 is a block diagram of the electrosurgical generator 102 of FIG. 1and a corresponding medical instrument 201 in accordance withembodiments of the present disclosure. The electrosurgical generator 102includes a controller 200, a high voltage power supply 202, and a radiofrequency output stage 204. The controller 200 includes a microprocessor206 and a memory 209. The microprocessor may be any suitablemicrocontroller, microprocessor (e.g., Harvard or Von Neumannarchitectures), PLD, PLA, or other suitable digital logic. Memory 209may be volatile, non-volatile, solid state, magnetic, or other suitablestorage memory.

Controller 200 may also include various circuitry (e.g., amplifiers orbuffers) that serves as an interface between the microprocessor 206 andother circuitry within the electrosurgical generator 102. Controller 200receives various feedback signals that are analyzed by themicroprocessor 206 to provide control signals based on the feedbacksignals. The control signals from controller 200 control the HVPS 202and the RF output stage 204 to provide electrosurgical energy to tissue,represented by a load 210 (Z _(load)).

The HVPS 202 includes an energy conversion circuit 208, which convertsAC from an AC source or direct current (DC) from a DC source at a firstenergy level into DC at a second different energy level. The energyconversion circuit 208 supplies the DC power at the second differentenergy level to the RF output stage 204 based on control signals fromthe controller 200. The RF output stage 204 inverts the DC power outputfrom the energy conversion circuit 208 to produce a high-frequencyalternating current (e.g., RF AC), which is applied to the load 210. Forexample, the RF output stage 204 may generate a high-frequencyalternating current using push-pull transistors coupled to a primaryside of a step-up transformer (not shown).

The electrosurgical generator 102 and controller 200 include circuitrythat determines and controls the power actually applied to the load 210(Z _(load)). The average power at the load 210 may be calculatedaccording to the equation:P _(avg) =V _(rms) ·I _(rm)·cos φ_(VI),where P_(avg) is the average power in watts, V_(rms) is theroot-mean-square value of the sinusoidal load voltage V_(load), I_(rms)is the root-mean-square value of the sinusoidal load current I_(load),and φ_(VI) is the phase angle between the load voltage V_(load) and theload current I_(load).

Alternatively, the average power may be calculated according to theequation:

${P_{avg} = {\frac{1}{T}{\int_{t_{1} - T}^{t_{1}}{{{v(t)} \cdot {i(t)}}\ d\; t}}}},$where T is the averaging time constant, v(t) is the load voltage as afunction of time, and i(t) is the load current as a function of time.The controller 200 uses the calculated average power at the load asfeedback to control the energy conversion circuit 208 so that theaverage power at the load is equal to a power level set by the user toachieve a desired tissue effect.

As shown in FIG. 2, electrosurgical generators typically include avoltage sensor 211 and a current sensor 212 coupled to the output of theRF output stage 204 to sense a voltage and a current for the averagepower calculations. The voltage sensor 211 measures the voltage acrossthe output leads of the RF output stage 204 and provides an analogsignal representing the measured voltage to an analog-to-digitalconverter (“ADC”) 215. ADC 215 converts the analog signal to a digitalsignal. The current sensor 212 measures the current on the output leadof the RF output stage 204 that is connected to the output terminal 104of the electrosurgical generator 102. The current sensor 212 provides ananalog signal representing the measured current to an ADC 215, whichconverts the analog signal to a digital signal.

In some electrosurgical generators, the digital voltage and currentsignals are used to calculate the average power at the load. However,processing delays associated with the measurement circuitry (i.e., thesensors 211, 212 and ADCs 215) and electrical parasitic components inthe cable 205 and in the measurement circuitry may introduce errors intothe voltage and current measurements. Because of errors in themeasurements, the magnitude of the measured voltage may not be equal tothe magnitude of the voltage actually applied to the load 210, and/orthe magnitude of the measured current may not be equal to the magnitudeof the current actually applied to the load 210, and/or the phasedifference between the measured voltage and current may not be equal tothe phase difference between the voltage and current actually applied tothe load 210. As a result, the average power calculated based on themagnitudes of the voltage and current and their phase difference may notbe equal to the average power actually applied to the load 210.

The systems and methods according to embodiments of the presentdisclosure minimize these measurement errors by introducing equalizersto equalize the power measurements made in the electrosurgical generator102 to the actual power applied to the load 210. As shown in FIG. 2, theelectrosurgical generator 102 incorporates equalizers 221 (e.g., filtersor algorithms). A first equalizer 221 is coupled in series with thevoltage sensor 211 and a corresponding ADC 215 and a second equalizer221 is coupled in series with the current sensor 212 and a correspondingADC 215.

The equalizers 221 are implemented in a digital signal processor (DSP)220 of the controller 200. The equalizers 221 receive measurements fromthe sensors 211, 212 and generate an estimated load voltage {circumflexover (V)}_(load) and an estimated load current Î_(load). The DSP 220also implements an average estimated power calculator 225 thatcalculates the average estimated power at the load {circumflex over(P)}_(avg) based on the estimated load voltage {circumflex over(V)}_(load) and the estimated load current Î_(load). The averageestimated power calculator 225 includes a multiplier 224 that multipliesthe estimated load voltage {circumflex over (V)}_(load) by the estimatedload current Î_(load) and an integrator 226 that integrates the outputfrom the multiplier 224 to obtain the average estimated power at theload {circumflex over (P)}_(avg).

The DSP 220 communicates the calculated average estimated power at theload {circumflex over (P)}_(avg) to the microprocessor 206, which usesthe average estimated power at the load {circumflex over (P)}_(avg) tocontrol the energy conversion circuit 208. For example, themicroprocessor 206 may execute a Proportional-Integral-Derivative (PID)control algorithm based on the average estimated power at the load{circumflex over (P)}_(avg) and a desired power level, which may beselected by a user, to determine the amount of electric current thatshould be supplied by the energy conversion circuit 208 to achieve andmaintain the desired power level.

FIG. 3 is a block diagram of an equalizer 221 that uses a least meanssquares (LMS) finite impulse response (FIR) adaptive filter according toan embodiment of the present disclosure. The equalizer 221 includes anLMS filter 302, an LMS weight adaptation unit 304, a desired responseinput unit 306, and an average estimated power ({circumflex over(P)}_(avg)) calculator 308. The equalizer may be implemented using apolyphase structure, such as the polyphase structure shown in FIG. 6B.The LMS filter 302 filters a digital input value x_(k) (e.g., a digitalvalue representing the measured voltage or the measured current) basedupon a weight vector W _(k+1) to produce a filtered output value y_(k).The weight vector W _(k+1) is produced by the LMS weight adaptation unit304 based upon the filtered output value y_(k) and a desired responsed_(k).

The desired response d_(k) for the LMS adaptation unit 304 may be apre-computed “pseudo-filter,” or time sequence. The desired responsed_(k) may have an idealized magnitude and phase versus frequencyresponse of a converged adaptive filter in the electrosurgical system.For instance, if the converged output current from the system matchesthe pre-measured magnitude and phase values at one or more frequencies,then this information is used to construct a sequence d_(k) and/or thepseudo-filter.

A desired response sequence d_(k) may be constructed during acalibration process for the electrosurgical generator 102. For a singlefrequency f₁, the calibration process first involves using the RF OutputStage 204 to generate the following test signal:x(t)=A ₁ sin(2πf ₁ t).where the amplitude A₁ is a measured or known value. The test signal isapplied to a resistive load (e.g., the test resistor 910 of FIG. 9),which is chosen to provide minimal phase shift for a nominal voltage andcurrent. The desired response unit 306 then generates a desired responsesequence d_(k).

The desired response sequence d_(k) is formed by sampling a sinusoidalcalibration signal d(t) having a known amplitude of excitation or thesame amplitude as the test signal x(t) (i.e., A₁), but delayed accordingto a measured or known phase θ₁ between the input of the ADCs 215 andthe output y_(k) of the adaptive filter (i.e., the combination of theLMS filter 302 and the LMS adaptation unit 304). In other words, thephase θ₁ represents the delays introduced by the ADCs 215 and otherelectronic or digital components disposed between the RF Output Stage204 and the output y_(k) of the LMS filter 302. Such a calibrationsignal may be expressed as follows:d(t)=A ₁ sin(2πf ₁ t+θ ₁).

For multiple frequencies f_(n), where n=1, . . . , N, the calibrationprocess involves using the RF Output Stage 204 to generate the followingseries of test signals:x _(n)(t)=A _(n) sin(2πf _(n) t),where n=1, . . . , N and the amplitudes A_(n) are measured or knownvalues. The series of test signals are summed together and applied to aresistive load (e.g., the test resistor 910 of FIG. 9).

The desired response sequence d_(k) for multiple frequencies is formedby sampling the sum of multiple sinusoidal calibration signals given bythe expression:d _(n)(t)=A _(n) sin(2πf _(n) t+θ _(n)),where n=1, . . . , N. The calibration signals d_(n)(t) have knownamplitudes of excitation or the same amplitudes as the respective testsignals x_(n)(t) (i.e., A_(n)), but are delayed according to measured orknown phases θ_(n) between the input of the ADCs 215 and the outputy_(k) of the adaptive filter (i.e., the combination of the LMS filter302 and the LMS adaptation unit 304).

At the end of adaptation, the estimated phases or delays of the voltageand current will be equal to or approximately equal to each other atdesired frequencies of interest, leaving only a difference between themeasured phases or delays of the voltage current. Also, the magnitudesof the measured voltage and current will be identical to orapproximately identical to the respective magnitudes of the estimatedvoltage and current.

FIG. 4 is a detailed block diagram of an equalizer 221 that uses an LMSfilter according to other embodiments of the present disclosure. The LMSfilter 302, which may be a finite impulse response (FIR) filter,includes a series of time shifting units 402 a-402 n and a series ofweighting units 404 a-404 n coupled to the digital input signal x_(k).During operation, the updated weight vector W _(k+1) is fed from the LMSadaptation unit 304 to the LMS filter 302 and becomes the current weightvector W _(k), which includes weight values w_(0k), w_(1k), . . . ,w_(Lk). The first weighting unit 404 a multiplies the digital inputsignal x_(k) by the first weight value w_(0k) of the current weightvector W _(k). The time-shifting units 402 b-n time shift the digitalinput signal x_(k) to obtain time-shifted digital input signals x_(k−1),x_(k−2), . . . , x_(k−L). The digital input signal x_(k) and thetime-shifted digital input signals x_(k−1), x_(k−1), . . . , x_(k−L)together form a digital input vector X _(k).

As shown in FIG. 4, the weighting units 404 b-n are connected torespective outputs of the time-shifting units 402 b-n. In thisconfiguration, the weighting units 404 b-n multiply the time-shifteddigital input signals x_(k−1), x_(k−1), . . . , x_(k−L) of the digitalinput vector X _(k) by respective weight values w_(1k), . . . , w_(Lk)of the current weight vector W _(k). The results of time-shifting andweighting the digital input signal x_(k) are added together by an adder406 to obtain the digital output signal y_(k).

The digital output signal y_(k) is fed back to the LMS weight adaptationunit 304, in which the digital output signal y_(k) is subtracted fromthe desired response d_(k) by a subtractor 408 to obtain a digital errorsignal e_(k). The LMS weight adaptation unit 304 includes an updatecomputation unit 410 that uses the digital error signal e_(k), the inputvector X _(k), and the weight vector W _(k) to compute an updated weightvector W _(k+1) according to the following LMS update equation:W _(k+1) =W _(k)+2μe _(k) X _(k),where μ is chosen by the designer and is bounded by:

$0 < \mu < {\frac{1}{\left( {L + 1} \right)\left( {{Signal}\mspace{14mu}{Power}\mspace{14mu}{of}\mspace{14mu}{\overset{\_}{X}}_{k}} \right)}.}$

The advantage of an equalizer 221 using the LMS filter 302 is that itcan accurately equalize the voltage and current measurements at allfrequencies of interest. The LMS filter may be trained when thegenerator is calibrated. The LMS filter 302 may also be trainedperiodically throughout the life of the electrosurgical generator 102.In some embodiments, once the LMS filter 302 is trained, the LMS weightadaptation unit 304 does not adapt the weight vector W _(k+1), but keepsit fixed.

FIG. 5 is an equalizer 221 according to another embodiment of thepresent disclosure. The equalizer 221 compensates for the gain and“phase” (in terms of delay) at a single frequency. The equalizer 221includes a gain unit 502 and a fractional delay line 504. The gain unit502 amplifies an input signal x(n) according to a gain correction factorK_(CF), which is determined from a calibration procedure describedbelow. During the calibration procedure, the gain correction factorK_(CF) is adjusted until the magnitude of the signal output from block504 matches a test signal (e.g., a measured or known reference signal)input to the voltage sensor 211 and the current sensor 212 of FIG. 2.This is similar to the results of the LMS adaptation at a singlefrequency described above.

The amplified input signal is then applied to the fractional delay line504, which may be expressed as z^(−Δt) ^(CF) , where Δt_(CF) is thetime-delay correction factor. The time-delay correction factor Δt_(CF)may be determined through a calibration procedure where the nominalphase or delay differences through the voltage sensor 211 and thecurrent sensor 212 are matched or made equal to a measured or knownreference value. The fractional delay line 504 may combine aninterpolation stage with a decimation stage to arrive at fractionalsample delay times. The fractional delay line 504 feeds an output signaly(n) to the average estimated power calculator 225 of FIG. 2.

The calibration procedure for determining the gain correction factorK_(CF) may involve using a test accessory 905 together with theelectrosurgical generator 102 of FIG. 2, as illustrated in the blockdiagram FIG. 9. The test accessory 905 includes a test resistance 910(R_(test)) that represents a load, a voltage reference meter 901 formeasuring the voltage across the test resistance 910, and a currentreference meter 902 for measuring the current passing through the testresistance 910. The test accessory 905 may include connectors that allowthe test accessory 905 to be removed from or connected to the terminals104, 106 of the electrosurgical generator 102. In other embodiments, thetest accessory 905 may be integrated into the electrosurgical generator102.

The test accessory 905 is used to calibrate the sensors 211, 212 andequalizers 221, 222 for magnitude and phase at one or more frequencies.The calibration process first involves applying the test resistanceR_(test) of the test accessory 905 across the output terminals 104, 106.The value of the test resistance R_(test) is selected to provide minimalphase shift for a nominal voltage and current. Then, the RF Output Stage204 generates one or more test signals at desired frequencies ω_(d).Next, a reference voltage magnitude ∥v∥ and a phase angle φ_(v) aremeasured at each of the desired frequencies ω_(d) using the voltagereference meter 201. Also, a reference current magnitude ∥i∥ and phaseangle φ_(i) are measured at each of the desired frequencies ω_(d) usingthe current reference meter 202. At the same time, the voltage sensor211, the current sensor 212, the ADCs 215, and the equalizers 221produce an estimated voltage magnitude ∥{circumflex over (v)}∥ and phaseangle {circumflex over (φ)}_(v) and an estimated current magnitude ∥î∥and phase angle {circumflex over (φ)}_(i) at each of the desiredfrequencies ω_(d) of the test signals.

For each desired frequency ω_(d), the gain correction factors for thevoltage and current equalizers K_(EQ) _(_) _(V)(ω_(d)) and K_(EQ) _(_)_(I)(ω_(d)) are calculated according to the following equations:

${K_{{EQ}\;\_\; V}\left( \omega_{d} \right)} = {{\frac{\hat{v}}{v}\left( \omega_{d} \right)\mspace{14mu}{and}\mspace{14mu}{K_{{{EQ}\;\_\; I}\;}\left( \omega_{d} \right)}} = {\frac{\hat{i}}{i}{\left( \omega_{d} \right).}}}$Then, for each desired frequency ω_(d), the minimum phases of theequalizers φ_(EQ) _(_) _(V)(ω_(d)) and φ_(EQ) _(_) _(I)(ω_(d)) aredetermined such that {circumflex over (φ)}_(v)={circumflex over (φ)}_(i)and φ_(v)−φ_(i)={circumflex over (φ)}_(v)−{circumflex over (φ)}_(i). Itis desirable to achieve “minimum” phase or delay because the voltage andcurrent measurements are in a closed loop and excessive phase or delayreduces the phase margin or bandwidth of the closed loop.

FIG. 10 is a flow diagram of a general method of calibrating anelectrosurgical generator according to embodiments of the presentdisclosure. After starting in step 1001, a resistive element havingappropriate characteristics is selected in step 1002. In step 1004, theresistive element is applied across the output terminals of theelectrosurgical generator. In step 1006, a test signal is generated at adesired frequency, and, in step 1008, the test signal is applied to theresistive element.

In step 1008, first magnitude values and first phase angle values ofvoltage and current components of the test signal are measured at theoutput terminals. In step 1010, second magnitude values and second phaseangle values for the voltage and current components of the test signalare estimated using a first equalizer for the voltage component (e.g.,the equalizer 221 of FIG. 2) and a second equalizer for the currentcomponent (e.g., the equalizer 221 of FIG. 2). In step 1012, gaincorrection factors, e.g., K_(EQ) _(_) _(V)(ω_(d)) and K_(EQ) _(_)_(I)(ω_(d)), for the first and second equalizers are determined basedupon the measured and estimated magnitudes of the voltage and currentcomponents of the test signal. Finally, before the calibration processends (step 1015), the minimum phase angle for the first and secondequalizers is determined in step 1014 based on the measured andestimated phase angles of the voltage and current components of the testsignal obtained in steps 1008 and 1010. The minimum phase angleinformation may be used to determine the time-delay correction factorΔt_(CF).

The fractional delay line 504 of FIG. 5 may be implemented with amulti-rate structure. FIG. 6A is a diagram of a multi-rate structure 600a for obtaining a fractional fixed delay of l/M samples. The inputsignal x(n), which has been sampled at the sample frequency F_(s), isapplied to an interpolator 602. The interpolator 602 up-samples theinput signal x(n) by a factor of M(F_(s)·L) to obtain an up-sampled orinterpolated signal v(m). The up-sampled signal v(m) is then filtered bya digital lowpass filter 604 to remove the images (i.e., the extracopies of the basic spectrum) created by the interpolator 602. Theresulting filtered signal u(m) is then delayed by l samples by a delayunit 606. Finally, the output w(n) from the delay unit 606 isdown-sampled by a factor of M by the decimator 608 to obtain anequalized output signal y(n) at the original sample frequency F_(s).

FIG. 6B is a diagram of an efficient polyphase implementation 600 b ofthe multi-rate structure of FIG. 6A. This implementation includes aseries of transversal FIR filters 612 a-612 n that filter the inputsignal x(n). The transversal FIR filters 612 a-612 n are given by thefollowing difference equation:p _(r)(n)=h _(Lowpass)(nM+r),0≤r≤(M−1).The delay of l is implemented as the initial position of the commutatorswitch (“l selector”) 614 corresponding to the sample at n=0.

FIGS. 7A-7C are diagrams of equalizers 221 that combine a simple gainwith an “all-pass” delay filter. The all-pass delay may be either afirst-order or second-order all-pass filter. The all-pass filter may bebetter at modeling the group delay across a relatively narrow bandwidthof interest than a simple gain combined with a fractional delay, whichmay be good at a single frequency, but may not be better than the LMSadaptive filter in magnitude and phase across a broad band offrequencies.

In the Laplacian s-domain, a first-order all-pass filter, which may beused to change delay or phase but not magnitude, is represented by thefollowing transfer function:

${A(s)} = {\frac{s - \alpha_{0}}{s + \alpha_{0}}.}$The magnitude of the first-order all-pass transfer function is:

${{A(s)}} = {{{H(s)}} = {\frac{{s - \alpha_{0}}}{{s + \alpha_{0}}} = {\frac{\sqrt{\alpha_{0}^{2} + \omega_{c}^{2}}}{\sqrt{\alpha_{0}^{2} + \omega_{c}^{2}}} = 1}}}$and the phase (in radians) is:

${{\beta\left( \omega_{c} \right)} = {{- 2}{\tan^{- 1}\left( \frac{\omega_{c}}{\alpha_{0}} \right)}}},$where the phase angle is 0 degrees when ω_(c)=0, −90 degrees whenω_(c)=α₀, and −180 degrees when ω_(c)>>α₀. By fixing ω_(c), the phaseβ(ω_(c)) is set by α₀. The group delay of the first-order all-passtransfer function is given by:

$T_{gd} = {\frac{2\alpha_{0}}{\alpha_{0}^{2} + \omega_{c}^{2}}.}$

The first-order all-pass filter is implemented in the digital domain.There are many ways to implement the first-order all-pass filter. Onemethod is to apply the bilinear transform by replacing the Laplacianvariable s with

${\frac{2}{T}\left( \frac{1 - z^{- 1}}{1 + z^{- 1}} \right)},$where T is the sample period. Then, the digital all-pass transferfunction becomes

${{H(z)} = \frac{1 - {k_{1}z^{- 1}}}{k_{1} - z^{- 1}}},{where}$$k_{1} = {\frac{1 + {\frac{T}{2}\alpha_{0}}}{1 - {\frac{T}{2}\alpha_{0}}}.}$This digital all-pass transfer function may be implemented by thefollowing difference equation:

${y(n)} = {{\frac{1}{k_{1}} \cdot {x(n)}} - {k_{1} \cdot {x\left( {n - 1} \right)}} + {\frac{1}{k_{1}} \cdot {y\left( {n - 1} \right)}}}$

Another method to implement a first-order all-pass filter is to use asimple feedforward/feedback digital comb filter having the followingtransfer function:

${{H(z)} = \frac{a - z^{- M}}{1 + {a \cdot z^{- M}}}},$where a is a constant and M is an arbitrary integer delay and M≥0. Thistransfer function may be implemented by the following differenceequation:y(n)=a·x(n)+x(n−M)−a·y(n−M).

FIG. 8 shows a digital circuit that implements the feedforward/feedbackdigital comb filter. The digital circuit includes first and secondadders 801, 803, first and second multipliers 802, 804, constant blocks811, 812, and a delay block 815, which provides a delay of M samples.The second multiplier 804 multiplies the output of the delay block 815by a constant −a. The first adder 801 adds the output from the secondmultiplier 804 to the input x(n) and provides the result to the delayblock 815. The first multiplier 802 multiplies the output from the firstadder 801 by a constant a. The second adder 802 adds the output of thefirst multiplier 802 to the output of the delay block 815 to produce theoutput y(n).

Another type of filter that combines an all-pass delay with a gain is ashelving filter. The shelving filter can be used to perform weighting ofcertain frequencies while passing other frequencies. The shelving filtercan be useful in emphasizing the signal band of interest. A first-orderparametric shelving filter transfer function is given by

${{H(s)} + {\frac{1}{2}\left\lbrack {1 + {A(s)}} \right\rbrack} + {\frac{V_{0}}{2}\left\lbrack {1 - {A(s)}} \right\rbrack}},$where A(s) is the first-order all-pass transfer function describedabove.

To implement a digital shelving filter, the first-order transferfunction H(s), which is in the s-domain, is converted to the z-domain.The transfer function H(s) may be converted to the z-domain using thebilinear transform to obtain the following transfer function:

${{H(z)} + {\frac{1}{2}\left\lbrack {1 + {A(z)}} \right\rbrack} + {\frac{V_{0}}{2}\left\lbrack {1 - {A(z)}} \right\rbrack}},{where}$${{A(z)} = \frac{- \left( {a + z^{- 1}} \right)}{1 + {a\; z^{- 1}}}},$

$a = \frac{{\tan\left( \frac{\omega_{c}T}{2} \right)} - V_{0}}{{\tan\left( \frac{\omega_{c}T}{2} \right)} + V_{0}}$for a frequency response that provides a cut, and

$a = \frac{{\tan\left( \frac{\omega_{c}T}{2} \right)} - 1}{{\tan\left( \frac{\omega_{c}T}{2} \right)} + 1}$for a frequency response that provides a boost.

The frequency response of the transfer function that provides a cutattenuates a range of frequencies and passes (i.e., applies a gain of 1to) an adjacent range of frequencies. On the other hand, the frequencyresponse of the transfer function that provides a boost amplifies arange of frequencies and passes an adjacent range of frequencies. Theresponse of the shelving filter may be modified by independentlycontrolling the cutoff/center frequency ω_(c) and the gain V₀.

The shelving filter transfer function H(z) may be implemented with theequalizer structures shown in FIGS. 7A-7C. As shown in FIG. 7A, theequalizer 221 includes an all-pass delay filter 711 (“A(z)”), an adder721, and a subtractor 722. The all-pass delay filter 711 filters theinput signal x(n) to obtain a filtered signal. The all-pass delay filterA(z) 711 may be implemented as a difference equation that is computedwith a digital signal processor. The adder 721 adds the filtered signalto the input signal x(n) and the subtractor 722 subtracts the filteredsignal from the input signal x(n).

The equalizer 221 of FIG. 7A also includes a first multiplier 723, asecond multiplier 724, and an adder 725 coupled together. The firstmultiplier 723 multiplies the output from the adder 721 by a first gain731 of 0.5 (or a 1-bit shift to the right) and the second multiplier 724multiplies the output from the subtractor 722 by a second gain 732 ofV₀/2, where V₀ is the gain of the all-pass filter when the frequency iszero. Finally, the adder 725 adds the outputs from the first multiplier723 and the second multiplier 724 to obtain the output signal y(n).

FIG. 7B is an equalizer 221 according to another embodiment of thepresent disclosure. The equalizer 221 of FIG. 7B includes the samecomponents and connections as the equalizer 221 of FIG. 7A except thatthe components and connections of the equalizer 221 of FIG. 7B arearranged differently. FIG. 7C is an equalizer 221 according to yetanother embodiment of the present disclosure. The input signal x(n) isfiltered by the all-pass delay filter 711 and then multiplied by thegain (1−V₀)/2 (733) using the first multiplier 723. The input signalx(n) is multiplied by the gain (1+V₀)/2 (734) using the secondmultiplier 724. Then, the adder 725 adds the results of the first andsecond multipliers together to obtain an equalized output signal y(n).

Another embodiment of the equalizer 221 may use a peak filter to boostor cut any desired frequency. A second-order peak filter may beimplemented with the equalizers 221 of FIGS. 7A-7C, where the all-passtransfer function in the Z-domain is given by:

${{A(z)} = \frac{z^{- 2} + {{d\left( {1 + a_{BC}} \right)}z^{- 1}} + a_{BC}}{1 + {{d\left( {1 + a_{BC}} \right)}z^{- 1}} + {a_{BC}z^{- 2}}}},{where}$${d = {- {\cos\left( \Omega_{C} \right)}}},{V_{0} = {H\left( e^{\Omega_{C}} \right)}},{a_{B} = \frac{1 - {\tan\left( \frac{\omega_{b}T}{2} \right)}}{1 + {\tan\left( \frac{\omega_{b}T}{2} \right)}}},{and}$$a_{C} = {\frac{V_{0} - {\tan\left( \frac{\omega_{b}T}{2} \right)}}{V_{0} + {\tan\left( \frac{\omega_{b}T}{2} \right)}}.}$The center frequency f_(c) of the peak filter is determined by theparameter d, the bandwidth f_(b) is determined by the parameters a_(B)and a_(C), and the gain is determined by the parameter V₀.

Using the equalizers 221 of FIGS. 7A-7C, power measurements may becalibrated in a manner similar to the examples described above by firstdetermining the desired gain and phase. The desired gain is determinedbased upon the difference between the measured ratio of gains and anideal or reference ratio of gains and the desired phase is determinedbased upon the difference between the measured phase and an ideal orreference phase. Then, it is determined whether the gain represents acut (e.g., V₀<0) or a boost (e.g., V₀>0). Finally, the digital all-passtransfer function A(z) having appropriate parameters is substituted intothe equalizers 221 of FIGS. 7A-7C.

Although the illustrative embodiments of the present disclosure havebeen described herein with reference to the accompanying drawings, it isto be understood that the disclosure is not limited to those preciseembodiments, and that various other changes and modifications may beeffected therein by one skilled in the art without departing from thescope or spirit of the disclosure.

What is claimed is:
 1. An electrosurgical generator comprising: anelectrical energy source; an output stage coupled to the electricalenergy source and configured to generate an electrosurgical signal basedon electrical energy output by the electrical energy source; a voltagedetector coupled to the output stage and configured to detect a voltageat an output of the output stage; a current detector coupled to theoutput stage and configured to detect a current at the output of theoutput stage; an equalizer configured to estimate voltage and currentapplied to a load based on the detected voltage and current; and a powercalculation unit configured to calculate power applied to the load basedon the estimated voltage and current, wherein the equalizer includes again element and an all-pass filter and is configured to minimize anerror between the calculated power and a power based on the detectedvoltage and current.
 2. The electrosurgical generator according to claim1, wherein the equalizer includes: a voltage equalizer configured toestimate the voltage applied to the load; and a current equalizerconfigured to estimate the current applied to the load.
 3. Theelectrosurgical generator according to claim 1, wherein the equalizer isa parametric equalizer.
 4. The electrosurgical generator according toclaim 3, wherein the parametric equalizer is one of a shelving boostfilter, a shelving cut filter, or a peak filter.
 5. The electrosurgicalgenerator according to claim 1, wherein the power calculation unit isconfigured to calculate actual power applied to the load.
 6. Theelectrosurgical generator according to claim 1, further comprisinganalog-to-digital converters electrically coupled to the current andvoltage detectors.
 7. The electrosurgical generator according to claim1, further comprising a digital signal processor, which includes theequalizer and the power calculation unit.